Multipath discriminator module for a navigation system

ABSTRACT

A multipath discriminator module for a communications and/or navigation system that implements spread spectrum modulation, which module has an input suitable for receiving navigation signals, a sampler for supplying sampled signals at a frequency twice the apparent frequency fc of the code of said signals, and a submodule for calculating an error signal e k  from said sampled signals, and a locally generated spreading code C|K|L wherein:  
         e   k     =       K   β                     ℜ        (           Z   K          --           Z   K     -       +     S                 β       )                 with                   {               Z   K   -     =       [       r     k   -   t2       ·     C          K           L         ]     ⊗     h   k   b                     Z   K   --     =       [       r     k   -   t1       ·     C          K           L         ]     ⊗     h   k   b                    
        or                   {                 Z   K   -     =       [       r   k     ·     C            K   -   t2             L         ]     ⊗     h   k   b                     Z   K   --     =       [       r   k     ·     C            K   -   t1             L         ]     ⊗     h   k   b                    
        with                   K   β       =       constant        
        with                   g        (   aTc   )         =       Sin                 Π                 α                 Cos                 Π                 β                 α       π                   α        [     1   -     (     2                 β                 a     )       ]                                   
 
     β designating the attenuation factor of the SRC signal.

[0001] The invention relates to a multipath discriminator module for anavigation system, and also to a navigation system including such amodule.

BACKGROUND OF THE INVENTION

[0002] Over the last ten years, direct sequence and spread spectrum(DS-SS) modulation systems have been increasing in importance.

[0003] At present, this technique is implemented not only in satellitenavigation systems such as GPS and GLONASS, but it has also beenintroduced into terrestrial and satellite communications systems, e.g.US standard IS-95, GLOBALSTAR, and more recently in the third generationof mobile telephones using the UMTS standard, and also in the Europeansatellite navigation system GALILEO.

[0004] The concept of DS-SS modulation, e.g. bi-binary phase shiftkeying (Bi-BPSK) introduces a pseudo-random noise (PRN) code which hasthe consequence of the resulting modulated signal presenting a passbandthat is wider than a signal that is transmitting only the data signals.It is in this sense that the spectrum density of the signal is said tobe “spread”.

[0005] In the receiver, a locally-generated replica of the transmittedPRN code is aligned with the phase of the code of the received signal.In particular, in navigation receivers, code phase alignment isessential for determining accurately the time of arrival (TOA), which isused for determining the geometrical distance between the transmitterand the receiver. Once alignment has been achieved, it is possible toestimate carrier phase and to determine the symbols of the transmitteddata.

[0006] This alignment is conventionally achieved in the receiver bymeans of a delay-locked loop (DLL), an example of which is described inthe article by M. Simon et al. published in the work “Spread spectrumcommunications handbook” published by McGrawHill, Inc., 2nd edition,1994.

[0007] Such alignment uses the result of correlation between thereceived signal and early and late versions (E and L) of alocally-generated reference code signal in order to calculate an errorsignal that is proportional to code phase error (the difference betweenthe estimated code phase and the received phase).

[0008] This error signal must indicate the direction in which the phaseof the reference signal needs to be offset (advanced or retarded) inorder to be brought into synchronization with the received signal. Thespacing between the early and late codes (E and L) is generally one bitof a pseudo-noise sequence known as a “chip”.

[0009] Signals of the square-root raised cosine (SRC) type (which have araised cosine spectrum) are defined in the UMTS standard. The same typeof SRC signal is likely also to be adopted in the above-mentionedGALILEO system. A digital receiver implementing such signals isdescribed in the article by R. de Gaudezni et al. entitled “A digitalchip timing recovery loop for band-limited direct-sequencespread-spectrum signals” published in IEEE Trans. on Comm., Vol. 41, No.11, pp. 1760-1769, November 1993.

[0010] The accuracy with which time of arrival is measured is negativelydisturbed by the presence of distortion due to multiple paths, and as aresult, when performing telemetry, the precision with which position isdetermined is decreased, and when transmitting data, there is anincrease in bit or frame error rate. This is particularly true when themultipath distortion is represented essentially by a single reflectioncoming from a point which is situated in the immediate environment ofthe receiver with a small dynamic range.

[0011] The superposition of the direct and reflected signals is thusliable to give rise to jitter which affects the TOA measurementsperformed by the DLL.

[0012] As a result, techniques that make it possible to reduce theimpact of multiple paths on determining code phase are of very greatinterest, particularly in the field of navigation.

[0013] Until now, methods for compensating multiple paths for use intelemetry have been developed essentially in the context of GPSreceivers.

[0014] As a result, a large number of those algorithms make use of thefact that the apparent chip rate of the publically available C/A code ismuch lower than the passband of the transmission. It is thenadvantageous to reduce the time differences between the early and latereference code signals E and L until they have a value that is less thanone bit of a pseudo-random noise sequence (“one chip duration”) in orderto reduce the error that is induced by the multipath beams.

[0015] In the context of SRC type systems, given that the frequencyspectrum is strictly limited to (1+β) times the apparent code rate(where β designates the attenuation factor of SRC pulses), theabove-mentioned methods are not effective in compensating multipathbeams.

[0016] In the article by Philip G. Mattos entitled “Multipathelimination for the low-cost consumer GPS” published in the Proceedingsof the ION GPS 1996 Conference in Kansas City, pp. 665-671, it has alsobeen suggested to replace the early and late correlation points E and Lby two early correlation points. However, that article does not give anymeans for implementing that technique.

OBJECTS AND SUMMARY OF THE INVENTION

[0017] An object of the present invention is to provide a discriminatormodule which is suitable for use in a spread spectrum communications ornavigation system, and more particularly one using SRC type modulation.

[0018] In a first variant, the invention provides a multipathdiscriminator module for a communications and/or navigation system thatimplements spread spectrum modulation, which module has an inputsuitable for receiving navigation signals, a sampler for supplyingsampled signals at a frequency twice the apparent frequency fc of thecode of said signals, and a submodule for calculating an error signale_(k) from said sampled signals, and a locally generated spreading codeC|K|L wherein:$e_{k} = {K_{\beta}\quad {\Re ( {\frac{Z_{K}\text{--}}{Z_{K} -} + {S\quad \beta}} )}}$${with}\quad \{ {\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\quad \{ {{\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}K_{\beta}} = {{{constant}{and}\quad S_{\beta}} = {{{- \frac{g( {- {t1Tc}} )}{g( {- {t2Tc}} )}}{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad \alpha \quad {Cos}\quad \Pi \quad \beta \quad \alpha}{\pi \quad {\alpha \lbrack {1 - ( {2\beta \quad a} )} \rbrack}}}}} } $

[0019] β designating the attenuation factor of the SRC signal.

[0020] In a second variant, the invention provides a multipathdiscriminator module for a communications and/or navigation system thatimplements spread spectrum modulation, which module has an inputsuitable for receiving navigation signals, a sampler for supplyingsampled signals at a frequency twice the apparent frequency fc of thecode of said signals, and a submodule for calculating an error signale_(k) from said sampled signals, and a locally generated spreading codeC|K|L wherein:$e_{k} = {K_{\beta}\quad ( {{- \frac{Z_{K}\text{--}}{Z_{K} -}} + {S\quad \beta}} )}$${with}\quad \{ {\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\quad \{ {{\begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}K_{\beta}} = {{{constant}{and}\quad S_{\beta}} = {{\frac{g( {- {t1Tc}} )}{g( {- {t2Tc}} )}{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad \alpha \quad {Cos}\quad \Pi \quad \beta \quad \alpha}{\pi \quad {\alpha \lbrack {1 - ( {2\beta \quad a} )} \rbrack}}}}} } $

[0021] β designating the attenuation factor of the SRC signal.

[0022] In a third variant, the invention provides a multipathdiscriminator module for a communications and/or navigation system thatimplements spread spectrum modulation, which module has an inputsuitable for receiving navigation signals, a sampler for supplyingsampled signals at a frequency twice the apparent frequency fc of thecode of said signals, and a submodule for calculating an error signale_(k) from said sampled signals, and a locally generated spreading codeC|K|L wherein:$e_{k} = {K_{\beta}\quad {( {{2\frac{Z_{K}\text{--}}{Z_{K}^{-} + Z_{K}^{+}}} + {S\quad \beta}} )}}$${with}\quad \{ {\begin{matrix}{Z_{K}^{+} = {\lfloor {r_{k + 0.5} \cdot C_{{K}L}} \rfloor \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k - {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{{kt} - {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\quad \{ {{\begin{matrix}{Z_{K}^{+} = {\lbrack {r_{k} \cdot C_{{{k + {1/2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{({k - {t2}})}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k} \cdot C_{{({k - {t1}})}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}\beta} = {{{constant}{and}\quad S_{\beta}} = {{{- \frac{g( {- {t1Tc}} )}{g( {- {t2Tc}} )}}{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad \alpha \quad {Cos}\quad \Pi \quad \beta \quad \alpha}{\pi \quad {\alpha \lbrack {1 - ( {2\beta \quad a} )} \rbrack}}}}} } $

[0023] β designating the attenuation factor of the SRC signal.

[0024] In each of the above cases, h_(k) ^(b) designates the impulseresponse of a lowpass filter.

[0025] The discriminator module may be such that:${K\quad \beta} = \frac{1}{\frac{}{ɛ}( \frac{g( {( {ɛ - {t1}} ){Tc}} )}{g( {( {ɛ - {t2}} ){Tc}} )} )_{ɛ = 0}}$

[0026] In each of the three above variants, it is possible to havet1=1.5 and t2=0.5.

[0027] The invention also provides a multipath discriminator module fora communications and/or navigation system that implements spreadspectrum modulation, which module has an input suitable for receivingnavigation signals, a sampler for supplying sampled signals at afrequency twice the apparent frequency fc of the code of said signals,and a submodule for calculating an error signal e_(k) from said sampledsignals, and a locally generated spreading code C|K|L wherein thesubmodule calculates the real portion of a ratio of two advancecorrelation values relative to the real phase value, these values comingfrom correlation between the received signal and the locally generatedreference signal.

[0028] The invention also provides a navigation system, presenting adiscriminator module as defined above.

[0029] Finally, the invention provides a navigation system, presenting adiscriminator module generating an error signal e′_(k) serving inconventional manner to correct a closed loop on the basis of sampledsignal Z+_(K) and Z−_(K) which is associated with a discriminator moduleas defined above in order to operate in an open loop on a said errorsignal e_(k) to generate a correction signal for the code phase output.

BRIEF DESCRIPTION OF THE DRAWINGS

[0030] Other characteristics and advantages of the invention appearbetter on reading the following description given by way of non-limitingexample and with reference to the accompanying drawings, in which:

[0031]FIG. 1 shows the influence of a multipath signal comprising adirect path and at least one reflected path;

[0032]FIG. 2 is a block diagram of a DLL as described in theabove-mentioned article by R. de Gaudenzi et al.;

[0033]FIGS. 3a and 3 b are graphs plotting the autocorrelation functiong(t) respectively for a module in accordance with FIG. 2 and inaccordance with the invention;

[0034]FIG. 4 is a block diagram of a DLL incorporating a discriminatorof the invention; and

[0035]FIG. 5 is a block diagram of a DLL incorporating a conventionaldiscriminator and a discriminator module of the present invention.

MORE DETAILED DESCRIPTION

[0036] Formula 3-1 represents a direct sequence spread spectrum (DS-SS)signal s_(T)(t) in baseband, with the spreading code word having alength of 2 pseudo-noise sequence bits (or “chips”), and each datasymbol dp,q,i presenting M/L distributed code words: $\begin{matrix}\begin{matrix}{{S_{T}(t)} = {\sqrt{\alpha \cdot P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}\quad {( {{{{d_{P}}_{\cdot}}_{{\lfloor i\rfloor}_{M}} \cdot c_{P \cdot {i}_{L}}} + {j \cdot b \cdot d_{Q \cdot {i}_{M}} \cdot c_{Q \cdot {i}_{L}}}} ) \cdot}}}} \\{{g_{T}( {t - {iT}_{c}} )}}\end{matrix} & ( {3\text{-}1} )\end{matrix}$

[0037] The factors a and b are as follows in the followingcircumstances:

[0038] a=1.0 and b=0 BPSK DS-SS

[0039] a=0.5 and b=1, with d_(p,|i|) _(M) =d_(Q,|i|) _(M) QPN DS-SS

[0040] a=0.5 and b=1, with d_(p,|i|) _(M) =d_(Q,|i|) _(M) Bi-BPSK DS-SS

[0041] with:

[0042] Ps: transmitted power

[0043] d_(P/Q,i) data symbols (d_(P/Q,i∈[−1.1]))

[0044] C_(P/Q,i) bit (“chip”) of a spreading code word of length L(C_(P/Q,i∈[−1.1]))

[0045] T_(c)=1/f_(c) duration of one bit of a pseudo-noise sequence (orduration of one “chip”)

[0046] g_(T)(t) pulse shape of a bit or “chip” e.g. SRC

[0047] M data symbol length in the duration of one bit or “chip”

[0048] |i|_(M) int(i/M)

[0049] |i|^(M) imodM

[0050] After transmission by a channel with added white Gaussian noise(AWGN) presenting symmetrical spectrum density N_(o/2), the filteredreceived signal r(t) is given by formula (3-2). $\begin{matrix}\begin{matrix}{{r(t)} = {{S_{T}( {t - \tau} )} \otimes {g_{r}(t)}}} \\{= {\sqrt{a \cdot P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}{( {{d_{P \cdot {\lfloor i\rfloor}_{M}} \cdot c_{P \cdot {i}_{L}}} + {j \cdot b \cdot d_{Q \cdot {\lfloor i\rfloor}_{M}} \cdot C_{Q \cdot {i}_{L}}}} ) \cdot}}}} \\{{{{g( {t - \tau - {iT}_{c}} )} \cdot {\exp ( {j( {{\Delta \quad \omega \quad (t)} + {\varphi \quad (t)}} )} )}} + {n(t)}}}\end{matrix} & ( {3\text{-}2} )\end{matrix}$

[0051] g(t)=g_(T)(t){circle over (x)} g_(T)(t) designating the pulseshape of one bit or “chip” after filtering and {circle over (x)}designating convolution.

[0052] g(t) thus constitutes the autocorrelation function of g_(T)(t).

[0053] Δω(t) designates the residual difference of the carrier frequencyafter the signal has been mixed into baseband.

[0054] φ(t) designates the phase of the carrier phase of the receivedsignal.

[0055] Because of the similarity between the in-phase and quadraturecomponents of the signal given by formula 3-2, it is appropriate toconserve only the in-phase component below. As a result, the symbols p,qdesignating these two situations are not conserved, in order to make thenotation more readable.

[0056] Formula 3-2 thus becomes: $\begin{matrix}\begin{matrix}{{r(t)} = {\sqrt{P_{s}} \cdot {\sum\limits_{i = {- \infty}}^{\infty}{d_{{\lfloor i\rfloor}_{M}} \cdot c_{{i}_{L}} \cdot {g( {t - \tau - {iT}_{c}} )} \cdot}}}} \\{{{\exp ( {j( {{\Delta \quad \omega \quad (t)} + {\varphi (t)}} )} )} + {n(t)}}}\end{matrix} & ( {3\text{-}3} )\end{matrix}$

[0057] for a system of the BPSK DS-SS type, for example.

[0058]FIG. 1 is a model of multiple paths that is suitable for use inshowing the effect of multiple paths on the DLL. In addition to thedirect signal coming from the satellite, the antenna also receives adelayed second version of the same signal, referred to as the multipathcomponent and due to reflection, with the delay in this second signalcoming from the fact that it has had to follow a path that is longer.

[0059] The sum of these two signals as received by the antenna can beexpressed using the following formula:

S′ _(T)(t)=S _(T)(t−τ)+α·S _(T)(t−τ−Δτ)·exp(jφ)  (3-4)

[0060] with

[0061] α: attenuation of the reflected signal relative to the directsignal

[0062] Δτ: delay of the reflected signal relative to the direct signal

[0063] φ=2π.Δτ.fc/c: the phase offset of the carrier of the reflectedsignal relative to the direct signal (c=speed of light, fc=carrierfrequency).

[0064] After filtering (see formulae 3-2 and 3-3), the following isobtained:

r(t)=r(t)+α·r(t−Δτ)·exp(jφ)  (3-5)

[0065] The explanation is given taking account of one reflection only.In practice, there are several components corresponding to multiplereflection paths which are all superposed on the direct signal.

[0066] The above-mentioned document by R. de Gaudenzi et al. uses a DLLof architecture outlined below, when describing FIG. 2.

[0067] It should be observed:

[0068] that the analog-to-digital converter could be located at adifferent location; and

[0069] that the time offset for aligning the local reference signal withthe received signal may be applied either to the local reference signalor to the received signal.

[0070] The baseband signal r(t) obtained by filtering using a filter ofcharacteristic G_(T)(f) is sampled at twice the bit or “chip” frequency,i.e. at 2fc. The samples that correspond to half-integer instants(k+0.5) Tc+τ are directed to the DLL, whereas the other samplescorresponding to “integer” instants kTc+τ are directed to instantaneouscorrelation and tracking of carrier phase and data demodulation (circuitNCO).

[0071] The samples corresponding to half-integer instants are given by:$\begin{matrix}\begin{matrix}{{r_{k} + {1\text{/}2}} = {{\sqrt{P_{S}} \cdot \exp}\quad {( {j\quad \varphi} ) \cdot {\sum\limits_{i = {- \infty}}^{\infty}{d_{{\lfloor i\rfloor}_{M}} \cdot c_{{i}_{L}} \cdot}}}}} \\{{g{( {( {ɛ_{k} + k + {1\text{/}2} - i} )T_{c}} ) \cdot {+ n_{k + {1/2}}}}}}\end{matrix} & ( {3\text{-}6} )\end{matrix}$

[0072] These samples r_(k+1/2) are directed along two branches. In theupper branch (FIG. 2), the samples are delayed by one bit or “chip” Tcprior to being multiplied by the k^(th) value of the spreading codeC_(|k|) _(L) as locally generated by the spreading code generator SCGEN.This multiplication is followed in each of the branches by a lowpassfilter H^(b)(z).

[0073] This produces the following samples: $\begin{matrix}{{Z_{k}^{+} = {\lbrack {r_{k} + {{{1/2} \cdot c}{k}_{L}}} \rbrack \otimes h_{k}^{b}}}{Z_{k}^{-} = {\lbrack {r_{k} - {{{1/2} \cdot c}{k}_{L}}} \rbrack \otimes h_{k}^{b}}}} & \text{(3-7)}\end{matrix}$

[0074] where:

[0075] h_(k) ^(b) designates the impulse response of the lowpass filter.The passband of this filter is limited in practice on the low side by:

[0076] the rate f_(s) (f_(s)=f_(c)/M) of data symbols, otherwise usefulenergy is lost; or

[0077] the dynamic range between the transmitter and the receiver, thedistance between the transmitter and the receiver is assumed to beconstant relative to the passband of H^(b)(z).

[0078] The error signal e_(k) is generated as follows:

e _(k=)|_(Zk) ⁻|²−|_(Zk) ⁺|²  (3-8)

[0079] In order to obtain an error signal that can be used directly bythe operational circuit NCO, the signal e_(k) is generally filtered byanother digital filter, the loop filter, whose transfer function isH^(d)(z). Given that e_(k) is independent of the data symbols, thecharacteristics of H^(d)(z) are determined mainly by the responsedesired of the DLL to the dynamic range between the transmitter and thereceiver, the phase estimat or loop needing to be capable of tracking alinear distance between the transmitter and the receiver that isincreasing or decreasing, without any residual tracking error.

[0080] The characteristic n(ε) indicates how the error signal e_(k)depends on the code phase error ε.

n(ε)=E[e _(k)|ε_(k) =ε∀k]  (3.9)

[0081] E [•] designating probability.

[0082] This gives:

n(ε)=g ²[(ε−0.5)T _(c) ]−g ²[(ε+0.5)T _(c)]  (3.10)

[0083] with SRC coding having an attenuation factor β, g(εT_(c))becomes:${g( {ɛ\quad T_{c}} )} = {\frac{\sin ( {\Pi \quad ɛ} )}{\pi \quad ɛ} \cdot \frac{\cos ( {\pi \quad \beta \quad ɛ} )}{1 - ( {2\quad \beta \quad ɛ} )^{2}}}$

[0084] The autocorrelation function g(εT_(c)) with β=0.35 is shown inFIG. 3a. The early and late samples E and L of formula (3-8) are givenfor ε=0 (no code phase error).

[0085] According to formula (3-10), the resulting S-shaped curve isshown in FIG. 3a. Using this curve, the DLL controls the interpolator sothat the received signal is in alignment with the locally-generatedsignal (i.e. ε=0).

[0086] The discriminator of the invention (FIG. 4) implements two earlysamples E1(Z_(k) ⁻) and E2(Z_(k) ⁻⁻) at instants (e−t₁)T_(c) and(ε−t₂)T_(c), generated using the formula: $\begin{matrix}{e_{k} = {{\kappa_{\beta} \cdot {( {\frac{{Zk}^{--}}{{Zk} -} + S_{\beta}} )}\quad \cdot {(\bullet)}}\quad {designating}\quad {the}\quad {real}\quad {{portion}\quad \cdot K_{\beta}}\quad {is}\quad a\quad {{constant}\quad \cdot S_{\beta}}\quad {is}\quad {an}\quad {{offset}\quad \cdot {the}}\quad {two}\quad {samples}\quad {E2}\quad {and}\quad {E1}\quad {are}\quad {defined}}} & ( {{see}\quad ( {3 - 7} )} )\end{matrix}$

[0087] by:

Z _(k) ⁻ =└r _(k−t2) ·C _(P,|K|L) ┘{circle over (x)}h _(k) ^(b) et Z_(k) ⁻⁻ =└r _(k−t1) ·C _(P,|K|L) ┘{circle over (x)}h _(k) ^(b)

[0088] with, for example, t₁=1.5 and t₂=0.5.

[0089] It is also possible to generate these two samples by declaringthe local replica of the code using the formula:

Z _(K) ⁻ =└r _(k) ·C _(P,|(k−t) ₂ _()|L) ┘{circle over (x)}h _(k) ^(b)

and

Z _(K) ⁻⁻ =└r _(k) ·C _(P,|(k−t) ₁ _()|L) ┘{circle over (x)}h _(k) ^(b)

[0090] The offset should be selected so that the expected value of theerror signal E[e_(k)] is zero when the code phase error ε_(k) is zero.

[0091] Applying formula (3-12), this gives: $\begin{matrix}{E\lbrack { {{{( {\frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} + S_{\beta}} )} {ɛ_{k} = 0} \rbrack} \equiv 0}\Rightarrow{S\quad \beta}  = {{E\lbrack {{{- {( \frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} )}}ɛ_{k}} = 0} \rbrack} = {- \frac{g( {- {t1Tc}} )}{g( {- {t2Tc}} )}}}} } & ( {3 - 13} )\end{matrix}$

[0092] The slope factor K_(β) is preferably selected so that the valueof the slope:$E\lbrack {{\frac{\quad}{ɛ}e_{k}{ɛ_{k}}} = 0} \rbrack$

[0093] is equal to 1 when ε_(K)=0

[0094] This gives: $\begin{matrix}{K_{\beta} = {\frac{1}{E\lbrack {\frac{\quad}{ɛ}{( {\frac{Z_{\kappa}^{--}}{Z_{\kappa}^{-}} + S_{\beta}} )} {ɛ_{k} = 0} \rbrack} } = \frac{1}{{\frac{\quad}{ɛ}( \frac{g( {( {ɛ - t_{1}} ){Tc}} )}{g( {( {ɛ - {t2}} ){Tc}} )} )ɛ} = 0}}} & (3.14)\end{matrix}$

[0095] for example with t₁=1.5 and t₂=0.5.

[0096] As shown by formula (3-12), the discriminator e_(K) of theinvention is independent of the phase Φ of the carrier. This dependenceis eliminated by generating the ratio Z_(K) ⁻⁻/Z_(K) ⁻.

[0097] The autocorrelation function g(t) for β=0.35 is given in FIG. 3b.

[0098] The discriminator module can be used in two ways:

[0099] either it can be used directly to replace a known discriminator;

[0100] or else it can be integrated in a known discriminator in order toprovide open loop correction signals to reduce the induced multipatherror.

[0101]FIG. 4 is a block diagram of a DLL branch including adiscriminator module of the invention.

[0102] Comparing FIGS. 2 and 4, it can be seen that:

[0103] the branch which generated z_(k) in FIG. 2 is replaced by abranch which generates z_(k) ⁻⁻ with a delay element of duration 2T_(c);

[0104] the error signal e_(k) is calculated as a function of z_(k) inthe circuit DISCR using formula (3-12) corresponding to the module ofthe invention.

[0105] For a signal of SRC type, the S-shaped curve n(ε) is given by:$\begin{matrix}{{{n(ɛ)}{{\kappa\beta} \cdot {( {\frac{g( {( {ɛ - t_{1}} )T_{c}} )}{g( {( {ɛ - t_{2}} )T_{c}} )} + S_{\beta}} )}}}{with}{{g( {ɛ\quad T_{c}} )} = {\frac{\sin ( {\pi \quad ɛ} )}{\pi \quad ɛ} \cdot \frac{\cos ( {\pi \quad \beta \quad ɛ} )}{1 - ( {2\quad \beta \quad ɛ} )^{2}}}}} & ( {3 - 15} )\end{matrix}$

[0106] with for example t1=1.5 and t2=0.5

[0107] The discriminator module corresponds to the desired behavior towithin a good approximation, i.e. the output of the S-shaped curve isproportional to the input e (e=K.ε) for −0.5ε≦0.5.

[0108] By selecting K_(β) in application of formula (3-14) K=1 and e=ε.

[0109] The discriminator module can be used to perform open loopestimation to correct the code phase output of a conventional module, asshown in FIG. 4.

[0110] Compared with the FIG. 4 module, there is an additional branchhaving a delay element 2Tc in order to generate the signal Z_(K) ⁻⁻.

[0111] A digital filter H_(com) ^(d)(z) can be used as the lowpassfilter at the output from the new discriminator module.

[0112] The operation of the DLL branch remains unchanged compared to thecase shown in FIG. 2.

[0113] Code phase is corrected by the output from the new discriminatormodule which is fed through the lowpass filter H_(corr) ^(d)(z). Sincethe new module is less affected by multiple paths, the error containedin the code phase estimated in the DLL branch can be corrected to alarge extent.

[0114] In a variant, the formula can be replaced by an amplitudefunction:$e_{k} = {K_{\beta}( {\lambda - \frac{Z_{k}^{--}}{Z_{k}^{--}} + S_{\beta}} )}$

[0115] where λ is a non-zero number lying between −1 and +1.

[0116] In another variant, the error signal e_(k) may be determined by:$e_{k} = {K_{\beta} \cdot {( {\frac{2 \cdot z_{k}^{--}}{z_{k}^{-} + z_{k}^{+}} + S_{\beta}} )}}$

[0117] This expression is particularly suitable for open loop correction(FIG. 5). Because of the effect of averaging the outputs from thelowpass filters (Z_(k) ⁻+Z_(k) ⁺)/2, the noise power in the resultingvariable is effectively halved, thus leading to less noise in the signale_(k).

[0118] It should be observed that the sampling instants t₁=1.5 andt₂=0.5 can have other values, and the difference t₁−t₂ between thesesampling instants could be other than one bit or “chip”.

What is claimed is: 1/ A multipath discriminator module for acommunications and/or navigation system that implements spread spectrummodulation, which module has an input suitable for receiving navigationsignals, a sampler for supplying sampled signals at a frequency twicethe apparent frequency fc of the code of said signals, and a submodulefor calculating an error signal e_(k) from said sampled signals(r_(k+1/2)), and a locally generated spreading code C|K|L wherein:$e_{k} = {K_{\beta}{( {\frac{Z_{\kappa}--}{Z_{\kappa} -} + S_{\beta}} )}}$${with}\{ {\begin{matrix}{Z_{\kappa}^{-} = {\lbrack {r_{k \cdot {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{\kappa}^{--} = {\lbrack {r_{k \cdot {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix}{or}\{ \begin{matrix}{Z_{\kappa}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{\kappa}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } $

K_(β)=constant h_(k) ^(b) designating the impulse response of a lowpassfilter${{and}\quad S_{\beta}} = {- \frac{g( {- {t1Tc}} )}{g( {- {t2Tc}} )}}$${{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad a\quad {Cos}\quad {\Pi\beta}\quad a}{\pi \quad {a\lbrack {1 - ( {2{\beta a}} )} \rbrack}}$

β designating the attenuation factor of the SRC signal. 2/ A multipathdiscriminator module for a communications and/or navigation system thatimplements spread spectrum modulation, which module has an inputsuitable for receiving navigation signals, a sampler for supplyingsampled signals at a frequency twice the apparent frequency fc of thecode of said signals, and a submodule for calculating an error signale_(k) from said sampled signals (r_(k+1/2)), and a locally generatedspreading code C|K|L wherein:$e_{k} = {K_{\beta}( {\lambda - \frac{Z_{\kappa}--}{Z_{\kappa} -} + S_{\beta}} )}$

λ lying in the range −1 to +1 $\begin{matrix}{{with}\quad \{ \begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k - {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{k - {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } \\{{or}\quad \{ \begin{matrix}{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{{K - {t2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{k}^{--} = {\lbrack {r_{k} \cdot C_{{{K - {t1}}}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} }\end{matrix}$

K_(β)=constant h_(k) ^(b) designating the impulse response of a lowpassfilter $\begin{matrix}{{{and}\quad S_{\beta}} = {- \frac{g( {{- {t1}}\quad {Tc}} )}{g( {{- {t2}}\quad {Tc}} )}}} \\{{{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad a\quad {Cos}\quad \Pi \quad \beta \quad a}{\pi \quad {a\lbrack {1 - ( {2\quad \beta \quad a} )} \rbrack}}}\end{matrix}$

β designating the attenuation factor of the SRC signal. 3/ A multipathdiscriminator module for a communications and/or navigation system thatimplements spread spectrum modulation, which module has an inputsuitable for receiving navigation signals, a sampler for supplyingsampled signals at a frequency twice the apparent frequency fc of thecode of said signals, and a submodule for calculating an error signale_(k) from said sampled signals (r_(k+1/2)), and a locally generatedspreading code C|K|L wherein: $\begin{matrix}{e_{k} = {K_{\beta}\quad ( {{2\frac{Z_{K}^{--}}{Z_{K}^{-} + Z_{K}^{+}}} + S_{\beta}} )}} \\{{with}\quad \{ \begin{matrix}{Z_{K}^{+} = {\lfloor {r_{k + 0.5} \cdot C_{{K}L}} \rfloor \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k - {t2}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{--} = {\lbrack {r_{{k\quad t} - {t1}} \cdot C_{{K}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} } \\{{or}\quad \{ \begin{matrix}{Z_{K}^{+} = {\lbrack {r_{k} \cdot C_{{{k + {1/2}}}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{K}^{-} = {\lbrack {r_{k} \cdot C_{{({k - {t2}})}L}} \rbrack \otimes h_{k}^{b}}} \\{Z_{k}^{--} = {\lbrack {r_{k} \cdot C_{{({k - {t1}})}L}} \rbrack \otimes h_{k}^{b}}}\end{matrix} }\end{matrix}$

β=constant h_(k) ^(b) designating the impulse response of a lowpassfilter $\begin{matrix}{{{and}\quad S_{\beta}} = {- \frac{g( {{- {t1}}\quad {Tc}} )}{g( {{- {t2}}\quad {Tc}} )}}} \\{{{with}\quad {g({aTc})}} = \frac{{Sin}\quad \Pi \quad a\quad {Cos}\quad \Pi \quad \beta \quad a}{\pi \quad {a\lbrack {1 - ( {2\quad \beta \quad a} )} \rbrack}}}\end{matrix}$

β designating the attenuation factor of the SRC signal. 4/ Adiscriminator module according to claim 1, wherein:${K\quad \beta} = \frac{1}{\frac{\quad}{ɛ}( \frac{g( {( {ɛ - {t1}} ){Tc}} )}{g( {( {ɛ - {t2}} ){Tc}} )} )_{ɛ = 0}}$

ε designating the code phase error. 5/ A discriminator module accordingto claim 1, wherein t1−t2=1. 6/ A discriminator module according toclaim 5, wherein t1=1.5 and t2=0.5. 7/ A multipath discriminator modulefor a communications and/or navigation system that implements spreadspectrum modulation, which module has an input suitable for receivingnavigation signals, a sampler for supplying sampled signals at afrequency twice the apparent frequency fc of the code of said signals,and a submodule for calculating an error signal e_(k) from said sampledsignals, and a locally generated spreading code C|K|L wherein thesubmodule calculates the real portion of a ratio of two advancecorrelation values relative to the real phase value, these values comingfrom correlation between the received signal and the locally generatedreference signal. 8/ A navigation system, presenting a discriminatormodule according to claim
 1. 9/ A navigation system, presenting adiscriminator module generating an error signal e′_(k) serving inconventional manner to correct a closed loop on the basis of sampledsignal Z+_(K) and Z−_(K) which is associated with a discriminator moduleaccording to claim 1 in order to generate in an open loop a correctionsignal for the code phase output.